| Title of Invention | METHOD AND APPARATUS FOR EFFICIENT SYNCHRONIZATION IN SPREAD SPECTRUM COMMUNICATIONS |
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| Abstract | A method for use in synchronizing a mobile station with a base station using a primary synchronization code and plural secondary synchronization codes transmitted by the base station, comprising: receiving the primary synchronization code and plural secondary synchronization codes transmitted by the base station; providing one of a set of Golay complementary sequences as synchronization code; providing plural other ones of the set of Golay complementary sequences with minimal cross-correlation with the primary synchronization code as the secondary synchronization codes; and obtaining a primary synchronization and a secondary synchronization with the base station using the provided primary and secondary synchronization codes. |
| Full Text | FORM 2 THE PATENTS ACT 1970 [39 OF 1970] COMPLETE SPECIFICATION [See Section 10] "METHOD AND APPARATUS FOR EFFICIENT SYNCHRONIZATION IN SPREAD SPECTRUM COMMUNICATIONS" TELEFONAKTIEBOLAGET LM ERICSSON [PUBL], a Swedish company, of S-126 25 Stockholm, Sweden, The following specification particularly describes the nature of the invention and the manner in which it is to be performed:- I FTF.TD OF THE INVENTION The present invention relates to telecommunications and more particularly to synchronizing transceivers in a direct sequence spread spectrum radio communications system. BACKGROUND AND SUMMARY OF THE INVENTION Modern communication systems, such as cellular and satellite radio systems, employ various modes of operation (analog, digital, and hybrid) and access techniques such as frequency division multiple access (FDMA), time division multiple access (TDMA*), code division multiple access (CDMA), and hybrids of these techniques. Digital cellular communications systems have expanded functionaliry for optimizing system capacity and supporting hierarchical cell structures, i.e., structures of macrocells, microcells, picocells, etc. The term "macrocell" generally refers to a cell having a size comparable to the sizes of cells in a conventional cellular telephone system (e.g., a radius of at least about 1 kilometer), and the terms "microcell" and "picoccll" generally refer to progressively smaller cells. For example, a microcell might cover a public indoor or outdoor area, e.g., a convention center or a busy street, and a picocell might cover an office corridor or a floor of a high-rise building. From a radio coverage perspective, macrocells, microcells, and picocells may be distinct from one another or may overlap one another to handle different traffic patterns or radio environments. An typical cellular mobile radiotelephone system includes one or more base stations (BSs) and multiple mobile stations (MSs). A base station generally includes a control and processing unic which is connected to a core network type node like a mobile switching center (MSQ which in turn is connected to the public switched telephone network (PSTN). General aspects of such cellular radiotelephone systems are known in the art. The base station handles a plurality of voice or data channels through a traffic channel transceiver which is controlled by the control and processing unit. Also, each base station includes a control channel transceiver which may be capable of handling more than one control channel also controlled by the control and processing unit. The control channel transceiver broadcasts control informadon over the control channel of the base stadon to mobiles tuned (or locked) onto that control channel. The mobile receives the informadon broadcast on a control channel at its voice and control channel transceiver. The mobile processing unit evaluates the received control channel information, which includes the characteristics of cells that are candidates for die mobile to lock on to, and determines on which cell die mobile should lock Advantageously, the received control channel informadon not only includes absolute information concerning the cell with which it is associated, but also contains relative information concerning other cells proximate to the cell with which the control channel is associated. In North America, a digital cellular radiotelephone system using TDMA is called the digital advanced mobile phone service (D-AMPS), some of the characterisdcs of which are specified in the TIA/EIA/1S-136 standard published by the Telecommunications Industry Association and Electronic Industries Association (TIA/EIA). Another digital communication system using direct sequence CDMA (DS-CDMA) is specified by the TTA/EIA/IS-95 standard, and a frequency hopping CDMA communication, system is specified by the EIA SP 3389 standard (PCS 1900). The PCS 1900 standard is an implementation of the GSM system, which is common outside North America, that has been introduced for personal communication services (PCS) systems. Several proposals for the next generation of digital cellular communication systems are currendy under discussion in various standards setting organizations, which include the International Telecommunications Union (ITU), the European Telecommunications Standards Institute (ETS1), and Japan's Associadon of Radio Industries and Businesses (ARIB). An example third operation standard currendy being proposed by ETSI is the Universal Mobile Telecommunications System (UMTS) Terrestrial Radio Access (UTRA). Besides transmitting voice information, the next generation systems can be expected to carry packet data and to inter-operate with packet data networks that are also usually designed and based on industry-wide data standards such as the open system interface (OSI) model or the transmission control protocol/Internet protocol (TCP/IP) stack. These standards have been developed, whether formally or dt facto, for rmny years, and the applications that use these protocols are readily available. In most of these digital communication systems, communication channels are implemented by frequency modulating radio carrier signals, which have frequencies near 800 megahertz (MHz), 900 MHz, 1800 MHz, and 1900 MHz. In TDMA systems and even to varying extents in CDMA systems, each radio channel is divided into a series of time slots, each of which contains a block of information from a user. The time slots are grouped into successive frames that each have a predetermined duration, and successive frames may be grouped into a succession of what are usually called superframes. The kind of access technique (e.g., TDMAor CDMA) used by a communication system affects how user information is represented in the slots and frames, but current access techniques all use a slot/frame structure. Time slots assigned to the same mobile user, which may not be consecutive time slots on the radio carrier, may be considered a logical channel assigned to the mobile user. During each time slot, a predetermined number of digital bits are transmitted according to the particular access technique (e.g., CDMA) used by the system. In addition to logical channels for voice or data traffic, cellular radio communication systems also provide logical channels for control messages, such as paging/access channels for call-setup messages exchanged by base and mobile stations and synchronization channels for broadcast messages used by mobile stations or other remote terminals for synchronizing their transceivers to the frame/slot/bit structures of the base stations. In general, the transmission bit rates of these different channels need not coincide and the lengths of the slots in the different channels need not be uniform. Moreover, third generation cellular communication systems being considered in Europe and Japan are asynchronous, meaning that the structure of one base station is not temporally related to the structure of another base station and that mobile does not know any of the structures in advance. In such digital communication systems, a receiving terminal must find die timing reference of a transmitting terminal before any information transfer can take place. For a communication system using DS-CDMA, finding the timing reference corresponds to finding the boundaries of downlink (e.g., BS-to-MS) chips, symbols, and frames. These are sometimes called downlink chip-, symbol-, and frame-synchronizations, respectively. In this context, a frame is simply a block of data that can be independently detected and decoded. Frame lengths in today's systems typically fall in the range of ten milliseconds (ms) to twenty ms. This search for BS timing may be called a "cell search," and it includes identification of BS-specific downlink scrambling codes that are features of current DS-CDMA communication systems. A mobile or other remote terminal typically receives a signal that is a superposition (sum) of attenuated, faded, and disturbed versions of the signal transmitted by a BS. The slot and frame boundaries in the received signal are unknown to the MS to begin with, as are any BS-specific scrambling codes. The mobile therefore must detect and identify one or more BSs in the noise-like (for DS-CDMA) received signal and to identify the scrambling code used. In order to help synchronize the remote terminal to the BS and identify the BS-specific scrambling code, some communication systems provide that each BS signal includes an unscrambled part, which may be called a synchronization channel (SCH), which the MS can lock onto and perform a "cell search." Even after downlink synchronization is achieved, synchronization also must be achieved for uplink communications (e.g., MS-to-BS). This is because of the rapidly changing radio interface between the base and mobile stations as well as "roundtrip" propagation delays to and from the base station. When the mobile desires to transmit to the base station, it sends an initial transmission over an uplink channel (e.g., a random access channel (RACH)). Because of the already-achieved downlink synchronization between the base and mobile stations, the uplink transmission is roughly or coarsely synchronized, e.g., typically within one time/access slot. For example, the base station may only allow uplink transmissions from mobiles during eight distinct time/access slots. The mobile knows generally when those time slots begin when it synchronized to the downlink signals being broadcast by the base station. The assumption therefore may be made that the mobile is out of synchronization no more than one such time/access slot. In a direct sequence spread spectrum system, such a time/access slot corresponds to a predetermined number of chips, e.g., 256 chips. In this example, the largest roundtrip propagation delay, i.e., the amount which the mobile is out of synchronization with the base station, is assumed to be 255 chips. To achieve the synchronization, the one station's transmission includes some known signal (i.e., known to the other station). In the downlink direction, the known signal is sometimes referred to as a synchronization sequence or code or a primary synchronization sequence or code. In the uplink direction, the known signal may be referred to as a preamble sequence. In the random access, the mobile station transmits the uplink random access burst. The random access burst consists of a preamble (or preamble sequence) part and a message part The base station RACH receiver correlates the received signal samples with the known preamble sequence. Once the maximum preamble correlation is detected by the base station RACH receiver, the base station is in synchronization and can accurately decode the substantive message from the mobile contained in the message part of RACH burst. Although only one random access synchronization code/preamble is needed for each base station, different synchronization codes/preambles associated with the base station may be used to minimize cross-correlation between uplink transmissions from different mobiles. In any event, the synchronization code(s)/preamble(s) used for the downlink cell search or for the uplink random access synchronization should have a minimal, out-of- phase, aperiodic autocorrelation. Autocorrelation is the property which describes how well a code or sequences correlates with itself. Minimal, out-of-phase, aperiodic autocorrelation means the autocorrelation values for the non-zero time shifts (i.e., the base and mobile stations are out of synchronization by one or more chips) are low compared to the zero-time shift autocorrelation value (the base and mobile stations are chip synchronized). Non- zero time shift autocorrelation values are called autocorrelation sidelobes. A zero time shift autocorrelation value is called the autocorrelation main lobe. One possible source for synchronization codes with unit envelope are binary or polyphase Barker codes. The main lobe to maximum sidelobe ratio for Barker codes is equal to L, where L is the code length. Unfortunately, binary Barker codes exist only for lengths 2,3, 4, 5, 7, 11, and 13, while polyphase Barker codes are currently known for the lengths up to 45. As a result, non-optimum codes have to be used when longer preamble synchronization codes are required. For example, the preamble of random access burst signal proposed for a third generation mobile communication standard (UTRA) has a length of 4096 code chips. Besides minimal aperiodic autocorrelation and longer code length, it is also important to efficiently generate and correlate the synchronization codes. With respect to correlation at the base station, efficiency is even more important if there are a number of different synchronization preambles/codes which can be randomly used so that the base station random access channel receiver must correlate a received composite signal with all possible synchronization/preamble codes. In this situation, the base station receiver employs a bank of random access preamble correlators, each of which performs a large number of data processing operations — especially if the code length is long. A similar problem exists with respect to the design and detection of the downlink synchronization codes cransmitted by the base station so that the mobile can identify the base station timing and obtain frame, symbol, and chip synchronization (i.e., cell search). To achieve efficiency, the proposed UTRA random access burst preamble is generated using a serial (or hierarchical) code concatenation procedure which has the benefit of reducing the number of correlation operations. The preamble is generated using a "signature" having 16 complex symbols. There are 16 different signatures obtained from an orthogonal set of binary orthogonal Gold sequences of length 16 by multiplying each binary sequence with the constant complex number C = (l +j)/√2, where j =√-1 Each signature symbol is spread with a so-called preamble spreading code, chosen to be a 256 chips long orthogonal Gold sequence. The resulting preamble sequence therefore has a length of 4096 chips, i.e., 16 x 256=4096. The preamble spreading code is cell specific and is broadcast along with information about allowed signatures in its cell by the base station. Unfortunately, the aperiodic autocorreladon properties of preambles based on serially concatenated orthogonal Gold sequences are not optimal, i.e., significant autocorrelation sidelobcs are generated. Significant autocorrelation sidelobes mean that the receiver may erroneously synchronize to one of these sidelobes, and as a result, the transmitted message is not properly received. It is therefore desirable to provide one or more synchronization sequences thac for longer lengths provide minimal aperiodic autocorrelation properties and which can be generated and conducted efficiendy. Such synchronization sequences could then be used advantageously in many types of spread spectrum communications applications such as downlink cell search and uplink random access. To meet these and other objectives, the present invention provides a correlator that may be included in a first transceiver used to correlate a received signal transmitted by a second transceiver. The correlator includes a matched filter corresponding to a complementary pair of sequences to correlate the received signal with one of the complementary pair of sequences. A detector detects a peak output from the matched filter, and timing circuitry uses the detected peak output to generate a timing estimate for synchronizing transmissions between the first and second transceivers. Each of the complementary sequences has very low, aperiodic autocorrelation sidelobe values for all non-zero delays of chat complementary sequence and a maximal autocorrelation main lobe value for a zero delay of thac complementary sequence. In one example application, the first transceiver may be a base station, and the second transceiver may be a mobile station. The one sequence may be used as a preamble portion of a random access burst transmitted by the mobile station to the base station over a random access channel. Alternatively, the first transceiver may be a mobile station, and the second transceiver may be a base station. The one sequence may be used as a synchronization sequence transmitted by the base station over a synchronization or other broadcast ape of channel. A plurality of such matched filters may be employed as a bank of correlators at the first transceiver with each matched filter corresponding to a complementary pair of sequences having minimal aperiodic autocorrelation sidelobe properties. This is particularly advantageous when there is a need for plural sequences from an orthogonal set to be used in the same cell, e.g., plural random access channels used, for example, to increase capacity. In a preferred example embodiment, the complementary pair of sequences is a complementary pair of binary sequences of length L = 2 , where -V is a positive integer. The complementary pairs of binary sequences are usually called Golay complementary pairs, and sequences forming such pairs are called Golay complementary sequences. The Golay complementary pair of sequences preferably has a relatively long length, e.g., L - 4096 chips. The present invention also provides for a sequence generator that efficiently generates a pair of Golay complementary sequences of length L = 2N that requires only N memory elements. A modulo 2N counter having N memory elements cyclically counts from zero to 2 . A permutator coupled to receive and permute N parallel outputs from the counter according to a certain permutation of integers ranging from one to N A first set of N logical AND operators produces N parallel outputs from pairs of adjacent permuted counter outputs. These outputs are then summed modulo 2 in a first summer. A second set of N logical AND operators operates on each permuted counter output and one of a set of Nweighting coefficients to produce N parallel outputs. A second summer sums modulo 2 the outputs generated by the second set of N logical AND operators. A third summer sums modulo 2 the outputs of the first and second summers to produce a first binary sequence in the complementary pair of binary sequences. A fourth summer sums modulo 2 the output of the third summer and a most significant output from the permutator to produce a second binary sequence in the complementary pair of binary sequences. In addition to an efficient synchronization sequence generator, the present invention also provides for an efficient sequence correlator for efficiendy correlating a received spread spectrum signal with a pair of binary (Golay) or polyphase complementary sequences of length L = 2' . Such a correlator is sometimes referred to as the efficient Golay correlator (EGC). The efficient sequence correlator includes N serially concatenated processing stages, each stage having first and second parallel processing branches. Each stage of the first processing branch includes a delay line coupled to a corresponding adder. Each stage of the second processing branch includes a multiplier coupled to a subtractor. The input signal provided to the first processing branch at the particular processing stage is stored in plural memory elements of the corresponding delay line, while the content of the last memory element of the same delay line is input to the adder (in the first branch) and to the subtractor (in the second branch) of die same stage. The input signal provided to the second processing branch at the same stage is multiplied in the multiplier by a corresponding weighting coefficient. The output of the multiplier is fed to the negative input of the subtractor (in the second branch) and to the adder (in the first branch). The output of the adder is then input to the delay line in the next successive srage. The output of the subtractor is then input to the multiplier in the next successive stage. The input signals for the first and the second processing branches at the first processing stage correspond to die received spread spectrum signal. The output of the first processing branch at the N-th processing stage produces a cross-correlation of the received spread spectrum signal with a first sequence from a complementary pair of sequences. The output of the second processing branch at the N-th processing stage produces a cross-correlation of the received spread spectrum signal with a second sequence from a complementary pair of sequences. In a preferred example embodiment, the complementary sequences are binary or Golay complementary sequences. The present invention also provides for a memory efficient Golay correlator with substantially reduced memory for correlating a received signal using one of the sequences from a pair of complementary sequences of L= 2 . The one sequence from the complementary pair is represented as a concatenation of two shorter, constituent complementary sequences of length X. Each constituent sequence is repeated L/X rimes and is modulated by +1 or -1 according to a signature sequence of length Y = L/X . A first correlator of length X receives the spread spectrum signal and generates an intermediate pair of correlation values corresponding to the constituent complementary sequences. A first selector alternately supplies one of the intermediate pair of correlation values during each of successive time windows based on a current value of an interleaving sequence of length Y = L/X whose elements belong to the set {0,l}. The order of concatenation of the constituent sequences is determined by die interleaving sequence. A multiplier multiplies che selected intermediate correlation values with the corresponding bits of the signature sequence to generate a multiplied correlation output. The multiplied correlation output is summed with a feedback output generated by a delay line having X memory elements with che resulting sum itself being fed back to the input of the delay line. A final correlation value corresponds to the resulting sum after Y successive time windows have occurred. In a preferred example embodiment, N = 12, L = 4096, X = 256, and Y - 16, and the first correlator is an efficient Golay correlator such as the one just described above. In yet another embodiment, this memory efficient correlator correlates the received spread spectrum signal with plural orthogonal sequences generated using plural signatures. Plural multipliers each multiply the selected intermediate correlation values with an element of a different signature sequence to generate a corresponding multiplied correlation output. Plural summers arc provided with each summer corresponding to one of the multipliers and summing the multiplied correlation output with the feedback outpuc of a corresponding delay line. After processing all the elements in the signature sequence, a final correlation value is generated for each of the orthogonal sequences. One example embodiment is disclosed in which sixteen signatures are used to generated sixteen orthogonal Golay sequences. Another example embodiment employs thirty-two signatures to generate thirty-two orthogonal Golay sequences. As a result of the present invention, a single synchronization sequence or an orthogonal set of synchronization sequences are provided which have excellent aperiodic autocorrelation properties for relatively long sequence lengths. Equally significant, the present invention provides for efficient generation of and correlation with one synchronization sequence or an orthogonal set of such synchronization sequences. These synchronization sequences can be advantageously employed in one example application to uplink synchronization over a random access channel and downlink synchronization over a broadcast synchronization channel. BRIEF DESCRIPTION OF THE DRAWINGS The foregoing and other objects, features, and advantages of the invention will be apparent from die following description of preferred embodiments as well as illustrated in the accompanying drawings in which reference characters refer to the same parts throughout. While individual functional blocks and components are shown in many of the figures, those skilled in the art will appreciate that these functions may be performed by individual hardware circuits, by a suitably programmed digital microprocessor, by an application specific integrated circuit (ASIC), and/or by one or more digital signaling processors (DSPs). Fig. 1 is a function block diagram of an example mobile communication system in which the present invention may be advantageously employed; Fig. 2 is a function block diagram of an example radio station transceiver in which the present invention may be advantageously employed; Fig. 3 is a function block diagram illustrating additional details of the baseband processing block in the receiver chain of the radio station transceiver shown in Fig. 2; Fig. 4 is a diagram of a recursive Golay sequence generator, Fig. 5 is a diagram of a direct Golay sequence generator; Fig. 6 is a diagram of a general type of correlator; Fig. 7 is a diagram of an efficient Golay correlator; Figs. 8A and 8B show a signaling format of random access bursts over a random access channel; Fig. 9 is a flowchart illustrating procedures for a random access routine performed by a mobile station; Fig. 10 is a flowchart illustrating a random access routine performed by a base station; Fig. 11 is a diagram of a particular random access channel preamble correlator based on concatenated sequences; Fig. 12 is a random access preamble correlator also based on concatenated sequences in which an example embodiment of the present invention is incorporated; Fig. 13 is a diagram showing a bank of random access preamble correlators in which an example embodiment of the present invention is employed; Fig. 14 is a block diagram of another example embodiment of the present invention implementing a factored, efficient, Golay correlator; Fig. 15 is a graph illustrating autocorrelation value against delay for a random access preamble correlator; and Fig. 16 is a graph illustrating autocorrelation value against delay for a random access preamble correlator implementing the example embodiment of the present invention. DESCRIPTION OF THE DRAWINGS In the following description, for purposes of explanation and not limitation, specific details are set forth, such as particular embodiments, procedures, techniques, etc., in order to provide a thorough understanding of the present invention. However, it will be apparent to one skilled in the art that the present invention may be practiced in other embodiments mat depart from these specific details. For example, while the present invention is sometimes described in the context of a mobile radio station using uplink spreading codes, the present invention is equally applicable to other radio stations, e.g., radio base stations, and indeed, to any spread spectrum communications system. In other instances, detailed descriptions of well-known methods, interfaces, devices, and signaling techniques are omitted so as not to obscure the description of the present invention with unnecessary detail. The present invention is described in the non-limiting, example context of a universal mobile telecommunications system (UMTS) 10 shown in Fig. 1. A representative, connection-oriented, external core network, shown as a cloud 12 may be for example the Public Switched Telephone Network (PSTN) and/or the Integrated Services Digital Network (ISDN). A representative, connectionless-oriented external core network shown as a cloud 14, may be for example the Internet. Both core networks are coupled to corresponding sen-ice nodes 16. The PSTN/ISDN connection-oriented network 12 is connected to a connection-oriented service node shown as a Mobile Switching Center (MSC) node 18 that provides circuit-switched services. In the existing GSM model, the MSC 18 is connected over an interface A to a Base Station Subsystem (BSS) 22 which in turn is connected to radio base station 23 over interface A'. The Internet connectionless-oriented network 14 is connected to a General Packet Radio Service (GPRS) node 20 tailored to provide packet-switched type services. Each of the core network service nodes 18 and 20 connects to a UMTS Radio Access Network (URAN) 24 over a radio access network (RAN) interface. URAN 24 includes one or more radio network controllers 26. Each RNC 26 is connected to a plurality of base stations (BS) 28 and to any other RNC's in the URAN 24. In the preferred embodiment, radio access is based upon wideband, Code Division Multiple Access (WCDMA) with individual radio channels allocated using CDMA spreading codes. WCDMA provides wide bandwidth for multimedia services and other high rate demands as well as robust features like diversity handoff and RAKE receivers to ensure high quality. Each mobile station 24 is assigned its own scrambling code in order for a base station 20 to identify transmissions from that particular mobile station as well as for the mobile station to identify transmissions from the base station intended for that mobile station from all of the other transmissions and noise present in die same area. A downlink (BS-to-MS) broadcast channel and an uplink (MS-to-BS) random access channel are shown. A CDMA radio station transceiver 30 in which the present invention may be employed is shown in Fig. 2 in function block format. Those skilled in the art will appreciate mat other radio transceiver functions used in CDMA transceivers not relevant to the present invention are not shown. In die transmit branch, information bits to be transmit are received by a spreader 32 which spreads diose information bits over the available frequency spectrum, (for wideband CDMA this frequency band could be for example 5 MHz, 10 MHz, 15 MHz or more), in accordance with a spreading code generated by a spreading code generator 40. Controller 44 determines which spreading code should be provided by code generator 40 to spreader 32. The spreading code provided by code generator 40 corresponds to a radio channel in a CDMA communications system. Because a very large number of code symbols (sometimes called "chips") may be used to code each information bit, (depending on die current data rate in a variable data rate system such as a WCDMA system), the spreading operation considerably increases die data rate thereby expanding the signal bandwidth. The spread signal is provided to a modulator 34 which modulates the spread signal onto an RF carrier. An oscillator 42 generates an appropriate radio frequency carrier at a frequency selected by die controller 44. The modulated RF signal is men filtered and amplified in RF processing block 36 before being transmitted over the radio interface by way of antenna 38. Similar but reverse operations are carried out in die receive branch of the transceiver 30. An RF signal is received by antenna 38 and filtered in RF processing block 150. The processed signal is then RF demodulated to extract the complex baseband signal from die RF carrier in a RF demodulator 48 using a suitable RF carrier signal provided by the oscillator 44, Usually, a quadrature down-converter is used in the RF demodulator in order to decompose the received signal into real (i) and imaginary (Q) quadrature components. In this way, the phase shift of the incoming signal due to the channel fading and/or phase asynchronism between the transmitter and receiver oscillators can be detected and compensated. The RF demodulated, complex base-band signal is correlated with a preamble sequence to obtain synchronizadon and then de-spread in the baseband receiver 46 in accordance with a de-spreading code. Reference is now made to Fig. 3 which shows the baseband receiver 46 in further detail. The synchronizadon preamble is a binary sequence, and the preamble matched filter/correlator 51 includes two idendcal binary preamble correlators, one (52) correladng the / branch of the received baseband signal and the other (53) correlating theQ branch of the same signal. The preamble sequence is incorporated in the preamble matched filter 51 while the de-spreading code is generated by die code generator 40. The received, complex, baseband preamble signal is correlated in the corresponding synchronizadon preamble correlator 51 in order to achieve exact synchronizadon with the incoming signal, and to obtain the initial values of the fading channel coefficients used and tracked later on in a baseband (RAKE) demodulator 57. The timing information is obtained in the peak detector 54 by comparing the absolute correlation value calculated from the real and imaginary correlators 52 and 53 with a threshold. The peak detector 54 provides the delays (relative to the receiver timing) of the various multipath components of die received signal. The timing unit 56 uses this information to compensate the mutual delays of multipath components before their combination and demodulation in the base-band (RAKE) demodulator. The complex correlation values corresponding to the detected correlation peaks are stored in initial channel estimator 55 and used as die initial fading channel coefficients to be canceled in the baseband demodulator 57. The present invention utilizes as one or more synchronization sequences from complementary pairs of sequences. A fundamental property of a complementary pair of sequences is chat the sum of their aperiodic autocorrelation functions equals zero for all non-zero time shifts. In the preferred example embodiment, the synchronization sequences are based on binary, i.e., Golay, complementary sequences. Advantageously, the Golay complementary sequences may be constructed for any length L = 2 , where N is any positive integer, and also for lengths 10 and 26, or for any combination of these lengths. In addition to having the possibility for long sequence lengths, many Golay complementary sequences have low aperiodic autocorrelation sidelobes. While the following text focuses on Golay complementary sequences for purposes of description only, many aspects of the invention encompass any sequence belonging to pairs of arbitrary complementary sequences. If the sequences are of length 1 = 2 , there is a general method to construct polyphase complementary pairs of sequences, where the Golay complementary pairs of sequences are a special binary case, i.e., biphase. That general construction is defined by the following recursive relation. (1) 15 an(k) and bn(k) are two complementary sequences of length 2 , 5(k) is the Kronecker delta function, k is an integer representing the time scale, n is the iteration number, Dn is a delay, Pn,n =1,2,...,N, is any permutation of numbers {0,1,2,..., N- l}, and Wn is an arbitrary complex number of unit magnitude. If Wn has values +1 and -1, the binary (Golay) complementary sequences are obcained. Taking all possible permutations Pn and all 2 different vectors Wn, (n = 1,2,..., N), it is possible to construct N12N different complementary pairs. However, the number of different Golay sequences of length 2 is N12 If the delays are not powers of two as defined in equation (1), and/or if all the weighting coefficients Wn do not have the same magnitude, the construction in equation (1) produces multilevel complementary pairs. However, many aspects of the present invention also apply to multilevel complementary pairs. A general recursive scheme for die simultaneous generation of both sequences aN[k) and bN(k) from the complementary pair is shown in Fig. 4. While the recursive Golay sequence generator 60 shown in Fig. 4 produces die pair of complementary Golay sequences aN[k) and bN(k) as defined in equation (1), it requires a large number of memoryelements, especially for longer codes. For example, for sequences of length L=4096 and N=12, twelve delay lines (D, - Dl2) are required, where die first delay line has D1 memory elements, the second delay line has D2 memory elements, and so on. The total number of memory elements is therefore equal to L — 1 = 2' - 1, because according to equation (1) Dl+D2+....+DN = l + 2 + 4+...+2N-1=2N-l. A more efficient (and therefore preferable but not necessary) means to generate Golay complementary sequences is now described. The direct Golay complementary sequence generator illustrated in Fig. 5 produces exactly the same pair of Golay complementary sequences as me recursive Golay sequence generator 60 shown in Fig. 4. The direct Golay sequence generator 70 includes a modulo 2 counter 72 which includes only N memory elements which permit cyclic counting from zero to 2' - 1. The counter 72 parallel outputs correspond to the binary N representation of the positive integer variable k, where k = ∑ Bt (k)2t-1 ~ . A permutator 74 is coupled to receive N parallel outputs B1(k) ... BN(k) and permute those N parallel outputs according to a certain permutation of integers ranging from one toN. The permutadon vector employed in the permutator 74 is obtained by adding 1 to all numbers in the permutadon used in equadon (1). A first set of N - 1 AND gates 76 produces N - 1 parallel outputs from pairs of adjacent outputs from the permutator 74. A first summer 80 sums modulo 2 the outputs generated by the first set of AND gates 76. A second set of AND gates 78 logically AND each output from the permutator 74 with one of a set of N weighting coefficients W, ... W/V which are obtained by mapping 1 —> 0 and — 1 —» 1 using the binary vectors W„ employed in equation (1). A second summer 82 sums modulo 2 to the outputs generated by the second set of AND gates 78. A third summer 84 sums modulo 2 the outputs of the first and second summers to produce a first binary sequence in the complementary pair of binary sequences. A fourth summer 86 sums modulo 2 the output of the third summer 84 and the most significant output from the permutator 74, i.e., BP(N) (k) to produce a second binary sequence in the complementary pair of sequences. A mapper 88 transforms the first and second binary sequences having elements from the alphabet {0,1} into corresponding elements from the alphabet {-1,1} The direct Golay complementary sequence generator 70 is memory efficient and has fairly low complexity, e.g., complexity similar to a conventional linear feedback shift-register sequence (m-sequence) generator. These qualities make generator 70 particularly attracdve as a sequence generator in the random access burst transmitter of the mobile terminal and also in the base station transmitter. At the receiver, the received spread spectrum signal must be correlated with the transmitted Golay complementary sequences in order to detect the exact time when the complete sequence is received. Since die transmitted Golay complementary sequence is chosen to have minimal aperiodic autocorrelation, the likelihood that the detected maximum correlation value corresponds to the autocorrelation main lobe is very high. A general, straightforward implementation of a correlator 90 matched to the arbitrary synchronization sequence is illustrated in Fig. 6. The received signal r(k) is provided to a delay line having L - 1 memory elements. Each input signal and each memory element stage provides an input to L multipliers for correlation with the sequence represented by a[L- 1], a(L — 2), a[L— 3], ... a(0). Th e outputs of the multipliers are summed to provide the correlation output signal R(Τ) . The Golay correlator 90, while satisfactory to perform the correlation, is quite cosdy in terms of memory and data processing. For example, if L = 4096, 4095 memory elements, 4096 multipliers, and 4095 summers are required. Accordingly, a preferred embodiment of the present invention employs an efficient Golay correlator (EGC) 100 illustrated in Fig. 7. Functionally, the efficient Golay correlator 100 is a matched filter direcdy corresponding to the complementary sequences aN(k) and bN(k) defined by equation (1) What makes this correlator 100 particularly attractive is that it employs considerably less memory and fewer data processing operations than required with the general correlator 90 shown in Fig. 6. The efficient Golay correlator 100 performs the correlation of the input signal r(k) simultaneously with the two complementary sequences aN (k) and bN(k). The two matched filter correlator outputs Rra(τ) and Rrb(τ) arc thc corresponding aperiodic cross-correlation functions, i.e., values representing the degree to which the received signal r(£) matches the two sequences aN[k\ and bN{k\. The matched filter/correlator 100 uses the complex-conjugates of the weighting coefficients Wn defined in equation (1), denoted as Wn . The efficient Golay correlator 100 therefore correlates the received spread spectrum signal r(k) to the pair of Golay complementary sequences of length L = 2N using .V serially concatenated processing stages, each stage having first and second parallel processing branches. The input signal Rα(n-1) '(k) provided to the first processing branch at each processing stage is stored in the first memory element of the corresponding delay line Dn, (where n = 1,...,N), while the content of the last memory element of the same delay line Dn is input to the adder (in the first branch) and to the subtractor (in the second branch) of the same processing stage. The input signal Rb(n-1) ' (k) provided to the second processing branch at the same stage is multiplied in the multiplier by a corresponding weighting coefficient Wn , which is the complex conjugate of the weighting coefficient Wn defined for equation (1). The output of the multiplier is fed to the negative input of the subtractor and to the adder in the first branch. The output Rb(n-1) (k) of the adder is input to the delay line in the next successive stage. The output Rb(n) (k) of the subtractor is input to the multiplier in the next successive stage. The input signals Ra(O)(k) and Rb(O)(k) for the first and second processing branches at the first processing stage correspond to the received spread spectrum signal r{k). The output of the first processing branch at the N-tb. processing stage produces a cross-correlation Rra(k) of the received spread spectrum signal with the first sequence aN(k) of the Golay complementary pair of sequences. The output of the second processing branch at the N-th processing stage produces a cross-correlation Rrb(k) of the received spread spectrum signal with die second sequence bN [k) from die Golay complementary pair. The number of muldplicadons in the EGC is equal to log2(L). In contrast, a straightforward matched filter implementation of a Golay sequence correlator 90 shown in Fig. 6 would require a much larger number of muldplicadons corresponding to 2 • L. Note the straightforward implementation 90 requires two correlators to correlate a complementary pair while the efficient Golay correlator 100 simultaneously correlates a complementary pair. The number of additions in me efficient Golay correlator 100 is 2- log 2 (L). 'ess than in die straightforward matched filter implementation 90 where it would be 2 - (L - 1). The number of memory elements required for the efficient Golay correlator 100 is L — 1 but the general correlator 90 shown in Fig. 6 requires two sets (2(L- l)j of memory elements to correlate the pair of sequences. Thus, the efficient Golay correlator 100 saves IC space, reduces processing operations, saves power, and reduces cost. The present invention has particularly advantageous, non-limiting application in providing synchronization between a mobile station and a base station when the mobile station sends a message over a random access channel corresponding to the base station. One example format of such a random access channel is now described in conjunction with Figs. 8A and 8B. The random access channel format is based on a slotted ALOHA approach where a mobile station can make a transmission on the random access channel at a number of well defined time offsets relative to a particular frame boundary. The different time offsets are shown as access slots and are spaced 1.25 milliseconds apart. Information on what access slots are available in the current cell is broadcast by the base station on a downlink broadcast channel. The frame boundary is also defined by die broadcast channel of the current cell. The random access burst includes two portions: a preamble portion having a length of one millisecond and a message portion having a length often milliseconds. Between the preamble and the message, there may be an idle time period, e.g., 0.25 milliseconds, which allows for the detection (correlation) of the preamble and subsequent online processing of the message. A random access procedure performed by the mobile station is now described in conjunction wirh die random access (MS) routine (block 200) shown in Fig. 9. Initially, the mobile station must acquire synchronization to a particular base station cell (block 210). As described in the background, the mobile station performs this initial synchronization during a cell search for a base station to which it has the lowest path loss. The mobile station tunes to base station broadcast channel and processes the broadcast signal to detect a synchronization sequence. In one example embodiment, the synchronization is broken up into two steps. In the first step, the mobile detects a primary synchronization sequence which is common to all base stations. Once the strongest or maximum correlation is detected, the mobile station executes a second step where it correlates a secondary synchronization sequence particular to this base station. After acquiring synchronization to a particular cell, the mobile station acquires from the downlink broadcast channel information regarding the random access preamble spreading code and message scrambling code(s) used in this particular cell along with the information about available preamble signatures, available random access channel access slots, and other information (block 220). The mobile station then selects an access time slot and a signature from those that are available (block 230). Thereafter, the mobile station generates a random access burst using the signature and the preamble spreading code during the selected access time slot (block 240). The random access bursts transmitted by the mobile station must be received and properly correlated at the base station as described in conjunction with the random access (BS) routine (block 300) illustrated in flowchart format in Fig. 10. The base station receives a composite RF signal, converts that signal to baseband, and correlates the baseband signal with one or more preamble sequences (block 310). In particular, the base station employs a matched filter type correlator to correlate the baseband composite signal with the preamble sequence (block 320). The maximum correlation peak is then detected and used to estimate the frame, slot, and chip synchronization timing information. The peak detector (e.g., 54) provides the relative (with respect to the receiver riming) delavs of the various multipath components of die received signal (block 330). The estimated timing information is provided to a RAKE combiner (e.g., 57) for demodulating the message portion of the random access channel burst (block 340). The timing unit (e.g., 56) compensates the mutual delays of multipath components before their combination and demodulation in the RAKE demodulator (e.g., 57) The complex correlation values corresponding to the detected correlation peaks are used as the initial fading channel coefficients to be canceled in the RAKE demodulator. The random access channel (RACH) preamble correlator 400 illustrated in Fig. 11 is particularly efficient in terms of the amount of memory it uses. Efficiency is achieved by serially (hierarchically) concatenating a signature binary sequence and a preamble spreading code, i.e., each binary bit of the signature sequence is spread by the same preamble spreading code. The received signal is correlated in the matched filter 410 with the preamble spreading code having, for example, a length of Tmax = 256 chips. The intermediate correlation value produced by me matched filter 410 is then multiplied in multiplier 420 with an element of a signature sequence (provided by the base station over the broadcast channel) having for example a length of sixteen. A summer 430 sums the output of multiplier 420 with the output of a delay line 420 having the same number of elements as the length of the matched filter, e.g., 256. The output of the summer 430 is input to the delay line 440 as well as provided to a peak detector 450. The preamble correlator 400 operates as desired under the condition that the expected delay T of the received signal r(k) is limited to be |Τ| The correlation operation is synchronized with a rime window, e.g., 256 chips, during which the random access burst is expected to be received. The rime window length assumes that the mobile station has already attained downlink or primary synchronization with the base station. For each successive 256 chip window, the 256 correlation values generated by the matched filter 410 for that window are multiplied by one symbol of the sixteen symbol length signature sequence. The next symbol in the signature sequence is then selected at the beginning of the next 256 chip rime window. The multiplied correlation values are summed with die result of the previous summation performed at the same relative time position in the previous time window. The result pf the previous summation is obtained from die output of die delay line 440 and the result of that current summation thereafter is then stored in die delay line 440. The final correlation values output to the peak detector 450 is provided after sixteen time windows, i.e., sixteen repetitions of die summations at each time position widiin the 256 chip rime window. The full correlation peak is therefore equal to 4096 because in each iteration an intermediate correlation peak of value 256 is integrated. The peak detector 450 receives the correlation values both from / and Q receiver branches and calculates the absolute correlation value or squared absolute correlation value. The peak detector 450 is connected to the rest of the RACH receiver as shown for example in Fig. 3. The RACH preamble correlator 400 is particularly efficient because it requires considerably less memory than a conventional correlator. For example, if L = 4096, the RACH preamble correlator 400 requires just 511 memory elements, while the general correlator 90 shown in Fig. 6 would require 4095 memory elements. The reduced memory benefits of the structure of the preamble correlator 400 are advantageously employed to implement a memory efficient Golay sequence correlator 500 shown in Fig. 12 and having a structure similar to that shown in Fig. 11. Correlator 500 reduces die number of additions and multiplications required. The efficient Golay correlator 500 shown in Fig. 12 is based on the representation of a Golay sequence of length L = 2N = J T maxas a function of two shorter, constituent complementary sequences A(k) and B[k) of length Tmax. This function is a consequence of the general recursive construction in equation (1) of a complementary pair of sequences. Namely, if the general recursive construction starts from die arbitrary pair of complementary sequences, i.e., if the initial vectors a0(k) and bO(k) are taken to be a0(k)=A(k), bQ(k)=B(k),k= 0,1,2 Tmax-l, where A(k) and B(k) are the two complementary sequences of length Tmux the resulting pair of complementary sequences of length L = 2N = J . Tmax is generated after/ iterations. All delays Dn in equation (1) are multiplied by the length of constituent sequences (Tmax) The received signal r(k) is preferably received by an efficient Golay correlator 510 shown in Fig. 7, having for example a length L = 256 chips. The efficient Golay correlator 510 generates two cross-correlation values Rr (τ) and Rr (τ) corresponding to the Golay complementary pair of sequences. A switch or selector 520 receives the two correlation values. A control signal, referred to as an "'interleaving signal," controls the selector 520 to select either the Rr (r) correlation value if a "0" is provided or the RrB (r) correlation if a "1" is provided. Selector 520 thereby alternately supplies one of the intermediate pairs of correlation values during each of successive time windows based on the current value of the interleaving sequence. The order of concatenation of the shorter constituent sequences is determined by the interleaving sequence. The interleaving sequence in this example where the signature has length sixteen also has a length equal to sixteen. The output of the selector 520 is then multiplied in multiplier 420 by one of the symbols of the signature corresponding to the current 256 chip timing window in multiplier 420. The operations performed in multiplier 420, summer 430, delay line 440, and peak detector 450 are similar to that described with respect to Fig. 11. As in the case of the scheme from Fig. 11, the memory efficient Golay correlator 400 operates as desired under the condition that the expected delay T of the received signal r(k) is limited to be |τ| In cells where higher capacity is desirable, it is useful to construct a set of plural synchronization sequences. More synchronization sequences provide for die possibility of more capacity. Preferably, die sequences in such a set should be ordiogonal to minimize cross-correlation. The invention provides for die efficient generation of orthogonal sets of Golay sequences from the recursive construction of equation (1). A set of orthogonal Golay sequences can be generated from equation (1) for each permutation Pn , i.e., for each set of the delays Dn, by choosing an appropriate set of L/2 = 2*"' vectors W(v,n) of length N, i.e., v = 0,1 2N-1 - 1 and n = 1,2,3,...,N. Each vector W(v,n) produces a pair of complementary Golay sequences. As a result, a total of 2 sequences are provided in die orthogonal set of Golay sequences of length 2 . The set of 2N-l vectors W(v,n) of length N which produces from equation (1) die orthogonal set of Golay sequences may be generated, for example, in two different ways. A first algorithm is W(v,n) = (-1)BN-N+1(2V),v = 0,1 2N-1 - l,n = 1,2,3 N, (2a) where Bn (x) is the n-di bit in die N-bits long binary representation of some positive integer x, i.e., x=∑Bn(x)2n-1 (2b) The second algoridim is given by W(v,n) = (-l)BN-N+1(2v+,),v = 0,l,...,2'v-1-l, n= 1,2,3,...,N. Consider the following illustrative example. Assume that an orthogonal set of Golay sequences of length L = 8 = 2 where N = 3, is to be obtained using equation (1). Starting from any permutation P = {P0,P1 , P2} of numbers {0,1,2}, let P= {0,2,1}. The necessary weighting vectors W(v,n) of length 3 are obtained from equation (2a) and are given in the following Table 1. n 1 2 W(0,n) 1 1 1 W(\,n) 1 -1 1 W(2,n) -1 1 1 W(3,n) -1 -1 1 Table 1 The resulting orthogonal set 5(u,k) of Golay sequences of length 8 is given in Table 2. k 0 2 3 4 5 6 7 5(0,k) 1 ■ 1 -1 -1 5(1,K) -1 1 -1 1 -1 5(2,K) 1 -1 1 -1 5(3,K) -1 -1 -1 -1 1 5(4,K) -1 1 -I 1 -1 -1 5(5,k) -I -1 1 1 1 5(6,k) -1 1 -1 -1 -1 1 1 5(7,k) -1 1 -1 1 -1 1 Table 2 To obtain corresponding quadriphase codes, each Golay sequence may be multiplied with the constant complex number (l + j)/√2, where j = √- 1. A specific, advantageous application of orthogonal Golay sequences to RACH preambles is now described in conjunction with the memory efficient Golay correlator illustrated in Fig. 13. Fig. 13 shows a preferred example embodiment of the invention implementing a RACH preamble correlator 600 where more than one preamble sequence may be used. Each of the thirty-two correlators shown in the bank of correlators in Fig. 13 operates similarly to the correlator 500 shown in Fig. 12. If the constituent sequences are of length Tmax = 256, the permutadon vector Pn, and the weighting vectors Wn are of length 16, the resulting total length of each sequence in complementary pair is 4096. The resulting Golay sequences consist of sequences A(k) and B(k} repeated eight times, which are multiplexed by die selector 520 using the "interleaving" function which depends on the permutation vector Pn . The interleaving function I0 (k) is common for the first sixteen orthogonal preambles, and the interleaving function I1(k) is common for the remaining sixteen orthogonal preambles. Each preamble has a unique "signature" sequence, which depends on the weighting function Wn and on the permutation vector Pn . The interleaving functions select one or the other output of EGC according to the structure of the transmitted preamble. The first orthogonal set of 16 Golay sequences of length 4096, having the common constituent sequences A(k) and B(k) of length 256 (and a common interleaving function), are obtained by choosing a single permutation vector of length 16, along with 8 weighting vectors according to equation (2). All delays Dn in equation (1) are multiplied by the length of constituent sequences (TMAX = 256). The additional 16 orthogonal Golay sequences of length 4096 having the same constituent sequences A(K) and B(k), but interleaved by another selector 610 according to another interleaving functionI1[(k), are obtained by taking The permutation vector and the weighting vectors used in equation (1) to obtain the additional set of orthogonal Golay sequences are the same as for the first set of orthogonal Golay sequences. In cotal, there are 32 orthogonal Golay sequences of length 4096 having the common constituent sequences of length 256. More gcneraUy, it is possible to generate 27 such orthogonal Golay sequences, where J = L/Tmax , These interleaving and signature functions are illustrated by an orthogonal sec of Goby sequences of length 8 using the same example given in Tables 1 and 2 above. If Tmax = 2 and the constituent sequences arc A(K) = {l, l} and B(k) = {1,-1}, the set of sequences given in Table 2 may be represented as shown in Table 3. Table 3 The interleaving functions 1O(K) and I1(k) are I0(k)= {0,0,1,1} and I1(k) = {1,1,0,0} where '0' value selects the cross-correlation with the sequence A(k) and a value of '1' selects the cross-correlation with the sequence B(k). Hence, the interleaving function l1(k) is the reverted version of I0(k) The first four 'signature' functions are given below: SignatureO = {1, 1,1, -1}, Signaturel = {1,-1,1,1}, Signature2= {1, 1, -1, 1}, Signacure3 = {1,-1,-1,-1}. The values of the other four signature functions are the same, i.e., Signature4 = SignatureO, . Signature5=Signaturel, Siganture6=Signature 2, and Signature7=Signature3. The previous example application of the invention was to one or more correlators employed by the base station to receive and synchronize to random access bursts transmitted by mobile stations. Another example advantageous application of the invention mentioned earlier is to a cell search correlator used in the mobile station. During the initial cell search, the mobile station searches for the base station to which it has the lowest path loss. It then determines the time slot and frame synchronization of the base station, and the downlink scrambling code associated with the base station. The initial cell search is carried out in three steps: (1) Slot synchronization, (2) Frame synchronization and code-group identification, and (3) Scrambling-code identification. This other, non-limiting example application of the present invention is related to the first and second steps (1) and (2). The slot synchronization procedure in the mobile station uses a primary synchronization code broadcast by all base stations at the beginning of each time slot and which is common to all base stations. The synchronization procedure consists of correlating the incoming signal with the primary synchronization code. The output of the correlator will generate peaks for each ray from each base station within range of the mobile station. Detecting the position of the strongest peak gives the timing of the strongest base station "modulo" the time slot length. The slot synchronization is a critical step from the mobile station's point of view, because this operation is performed all the time, even if the mobile station already has a link to some other base station. Therefore, the implementation complexity of the mobile's correlator for the primary synchronization code should be minimized if possible. Such minimization may be achieved by exploiting a special structure of the primary synchronization code. On the other hand, the primary synchronization code should have minimum aperiodic autocorrelation sidelobes to minimize the probability of synchronizing the mobile station to a sidelobe rather than to the desired main lobe. This latter requirement for the cell search primary synchronization code may be fulfilled by using the Golay complementary sequences, as it is proposed in commonly-assigned U.S. patent application Serial No. 09/135,247 entided "Communication Methods and Apparatus Based on Orthogonal Hadamard-Based Sequences Having Selected Autocorrelation Properties," filed on August 17, 1998, the disclosure of which is incorporated by reference. That patent application does not specify an efficient method for the matched filtering of the primary synchronization code based on Golay complementary sequences for use in the first cell search step. Fortunately, the present invention provides an efficient implementation that uses Golay complementary sequences as primary synchronization codes. Using an efficient Golay correlator similar to EGC 100 shown in Fig. 7 as the primary synchronization code correlator in the first step of the cell search provides a larger implementation gain than, for example, a hierarchical correlator (HQ similar to the correlator 400 shown in Fig. 11 matched to a hierarchical primary synchronization code. The hierarchical code of length 256 is obtained by serial, i.e., hierarchical concatenation of two constituent sequences of length 16, as included in one proposal for the UTRA system. In the preferred embodiment, the length of primary synchronization code is 256 chips. The Golay complementary sequences of length 256 are optimum for application as primary cell search codes because they have a minimum maximum absolute aperiodic autocorrelation sidelobe (MAS) value. The minimum MAS value obtained for all Golay complementary sequences of length 256 is 12. One pair of the Golay sequences which has a MAS=12 is defined by Pn ={6,3,7,2,1,4,0,5}, and Wn = {1,-1,-1,1,1,-1,1,1}. (3) The number of memory bits Nmem for the EGC may be calculated by taking into account the number of memory cells in each delay line and die fact diat after the first delay line, the memory cell word length increases by 1 for each new delay line. Assuming a one-bit quantization of the received signal, it follows: The hierarchical correlator (HC) requires one straightforward correlator of length 16 and 15 delay lines of length 16. The memory cell word length for each such delay line is 5 bits. Therefore, Wmem(HC) = 1-16 + (5-16) 15= 1216 for the HC. The Golay complementary sequence correlated by the EGC offer the following advantages over the hierarchical sequences: • The number of adders is 16 for the EGC as compared to 30 adders for the HC. • The number of multipliers is 8 for the EGC as compared to 32 for die HC. • The number of memory bits is 30% less for the EGC than for the HC. • The maximum absolute value of aperiodic autocorrelation sidelobes is 3 times lower for Golay complementary sequences than for hierarchical sequences. Another example embodiment of die invention provides a factored efficient Golay correlator. Such a correlator of length 256 is shown in Fig. 14. A Golay sequence represented in the factored form can be constructed as a concatenation of two shorter constituent (complementary) sequences, e.g., of length 64- In that case, the correlator 700 includes an efficient Golay correlator 710 oflength 64, similar to diat shown in Fig. 7, delay lines 720, 730, 740, and 750 oflength 64, multipliers 790 and adders 760, 770, and 780. The total number of multipliers is at most 10, and the total number of adders is equal to 15. Note that die number of adders is 1 less than for the EGC oflength 256. The actual number of delay lines depends on the interleaving sequence. The correlator 700 may be useful when the coherence rime of the receiver local oscillator is shorter than the sequence length, (e.g., it is at most 64 chips). As a result, a sub-optimum, non-coherent integration of the intermediate correlation values is necessary. In other words, a correlator 700 allows both coherent and non-coherent integration of intermediate correlation values. The example correlator 700 in fig. 14 is coherent. In this example, the Golay sequence of length 256 is represented {-B, B, A, A}, where A and B are complementary sequences of length 64. In a second step of the cell search, a set of 17 Secondary Synchronization Codes (SSCs) of length 256 chips may be used. The SSCs should be orthogonal to the primar)- synchronization code (PSC). Assuming that the primary synchronization code is a Golay complementary sequence in accordance with one example application of the invention, one way to generate the set of 17 orthogonal SSCs is proposed in commonly-assigned U.S. patent application Serial No. 09/135,247 entided "Communication Methods and Apparatus Based on Orthogonal Hadamard-Based Sequences Having Selected Autocorrelation Properties," filed on August 17,1998, the disclosure of which is incorporated by reference. The orthogonal set of SSCs is generated by multiplying die PSC by 17 different Walsh functions of the same length. Such a set of SSCs allows the use of a Fast Hadamard Transform (FHT) processor in the second step of cell search in order to reduce the implementation complexity. Moreover, if the PSC is a Golay complementary sequence, the set of sequences obtained by multiplying that Golay PSC by different Walsh functions results in an orthogonal set of Golay sequences. In this case, these SSCs may also be detected using a bank of EGCs similar to die primary synchronization code correlation procedure described above. Such a bank of EGC in accordance with the present invention is more efficient in rermS of the number of required operations than correlation based on a FHT. An additional requirement for SSCs is diat the aperiodic cross-correlation between the PSC and each of 17 SSCs be small in order to minimize the interference from the secondary synchronization codes to the primary synchronization code correlator. In that case, it is advantageous to generate a complete set of 256 orthogonal Golay sequences of length 256 chips using equation (2), select one of these Golay sequences as the primary synchronization code, and then select 17 second synchronization codes that each have a maximum absoluce aperiodic cross-correlation with the primary synchronization code which is less than some reasonable direshold. For example, assuming that the PSC is an a(k) sequence generated by equations (1) and (3), the set of 17 SSCs having a maximum absolute aperiodic cross-correlation with PSC less than or equal to a threshold of 42 is defined by die same permutation vector as in equation (3) and by the following weighting vectors: (4) The corresponding sequence SSCn is equal to the Golay sequence a(k) or b(k) (indicated in the brackets attached to each vector in (4)) when the weighting vector WSSCn is replaced in (1). The SSCs can be detected by using the corresponding bank of 17 (or less) EGCs, similarly as for PSC. Besides the improved implementation efficiency, synchronization codes based on Golay complementary pair sequences offer superior performance in terms of the maximum absolute aperiodic autocorrelation sideiobes (MAS) as compared for example to die synchronization sequences based on concatenated Orthogonal Gold sequences. This comparison is illustrated by the graphs in Figs. 15 and 16. Fig. 15 is a graph of die aperiodic autocorrelation function of serially concatenated orthogonal Gold sequences. Note that although there is a substantial maximum at the sample corresponding to zero delay, diere are also significant maximum peaks which could cause false correlation detections at various non-zero delays. In contrast, the graph in Fig. 16 shows diat for synchronization codes based on Golay complementary pair sequences, the sideiobes are much lower, and in the range of interest for the random access detection, (i.e., +/- 255 chips around die main lobe), the sideiobes are even lower dian elsewhere. For the application of orthogonal Golay complementary sequences as the random access preambles, it is of interest to compare their aperiodic autocorrelation properties with those of anodier set of orthogonal preambles obcained by serial concatenation of orthogonal signatures with a common preamble spreading code. The parameter for comparison is the maximum absolute aperiodic autocorrelation sidelobe (MAS) of each preamble in the set. In diis example comparison which is directed to the currendy evolving UTRA system, a set of 16 orthogonal Golay sequences of length 4096 in accordance with the present invention is compared with anodier UTRA proposal for die RACH preambles based on orthogonal Gold sequences. Table 4 provides sixteen Golay sequences as a function of two shorter, constituent complementary sequences A(k) and Table 4 The constituent Golay sequences A(k) and B(k) are obtained from equation (1) using the permutation Pn ={0,2,1,5,6,4,7,3}, and weighting Wn={1,-1,1,-1,1,-1,-1,1}. The MAS values for the set of 16 orthogonal Golay sequences defined above, and for the set of 16 concatenated orthogonal Gold sequences obtained from 16 orthogonal signatures spread by a common preamble spreading code as described in another proposal for UTRA (the preamble spreading code number n = 1), are shown in Table 5. Golay sequences Concatenated Orthogonal Gold sequences Number of occurrences MAS Number of occurrences MAS 4 161 1 1024 4 181 4 1280 8 183 7 1536 - - 2 1792 - - 2 2048 Table 5 As described earlier, the random access preambles are not completely asynchronous to the base station receiver because the mobile terminal already has the basic information about base station timing, but there is an uncertainty introduced by the round-trip propagation delay between the base station and the mobile terminal. One reasonable assumption is that the round-trip delay is at most 255 chips, so the aperiodic autocorreladon function of random access preambles is actually of interest only in the region +/- 255 chips around the main lobe. The maximum absolute values of aperiodic autocorrelation sidelobes in the region +/- 255 chips around the main lobe are shown in Table 6 for the previously described orthogonal Golay and concatenated orthogonal Gold sequences of length 4096. Golay sequences Concatenated Orthogonal Gold sequences Number of occurrences MAS Number of occurrences MAS 16 31 1 731 - - 2 737 - - 3 743 - - 3 755 - - 6 761 - - 1 767 Table 6 Table 6 shows that the orthogonal Golay sequences have about 25 times lower autocorrelation sidelobes than the concatenated orthogonal Gold sequences, in the region +/- 2S5 chips around the main lobe. While the present invention has been described with respect to a particular embodiment, those skilled in the art will recognize that the present invendon is not limited to the specific embodiments described and illustrated herein. Different formats, embodiments, and adaptations besides those shown and described as well as many modifications, variations, and equivalent arrangements may also be used to implement the invention. Therefore, while the present invention has been described in relation to its preferred embodiments, it is to be understood diat this disclosure is only illustrative, in no way restrictive, and merely exemplary of die present invention. WE CLAIM : 1. A method for use in synchronizing a mobile station with a base station using a primary synchronization code and plural secondary synchronization codes transmitted by the base station, comprising: receiving the primary synchronization code and plural secondary synchronization codes transmitted by the base station; providing one of a set of Golay complementary sequences as synchronization code; providing plural other ones of the set of Golay complementary sequences with minimal cross-correlation with the primary synchronization code as the secondary synchronization codes; and obtaining a primary synchronization and a secondary synchronization with the base station using the provided primary and secondary synchronization codes. 2. The method as claimed in claim 1, wherein the primary synchronization code and 17 secondary synchronization codes are selected from a set of 256 orthogonal Golay sequences. 3. The method as claimed in claim 1, wherein the minimal cross-correlation is determined using a threshold. 4. The method as claimed in claim 1, wherein the minimal cross-correlation is minimal aperiodic cross-correlation. 5. The method as claimed in claim 1, wherein the synchronization circuitry includes a bank of Golay correlators to correlate a signal received by the mobile station with the primary and secondary synchronization codes. 6. The method as claimed in claim 1, wherein the set of Golay complementary sequences has a property that the sum of their aperiodic autocorrelations equals zero for all non-zero shifts, the method further comprising: using a matched filter corresponding to a pair of Golay complementary sequences to correlate a received signal with at least one of the Golay complementary pair of sequences; detecting a peak output from the matched filter; using the detected peak output to generate a timing estimate for synchronizing transmissions between the mobile station and the base station. 7. The method as claimed in claim 6, further comprising: using complex correlation values corresponding to detected peak outputs as initial fading channel coefficients employed to demodulate a message include in the received signal. 8. The method as claimed in claim 6, wherein the matched filter correlates the received signal with both Golay complementary sequences at the same time. 9. The method as claimed in claim 6, wherein the matched filter correlates the received signal with the complementary sequences recursively. 10. The method as claimed in claim 1, wherein at least one of the Golay complementary sequences is included in a preamble portion of a random access message transmitted by the mobile station to the base station over a random access channel. 11. The method as claimed in claim 1, wherein at least one of the Golay complementary sequences is transmitted by the base station over a broadcast channel to permit the mobile station to synchronize at least roughly with the base station. 12. The method as claimed in claim 1, wherein each Golay complementary sequence is generated by concatenating a pair of shorter Golay sequences. 13. The method as claimed in claim 1, wherein a complementary pair of sequences is generated in the sequence generation circuitry in accordance with the following: ao (k) = 5( k) bo(k)= S (k) an(k) = an-i(k) + Wn ■ bn-1 (k -Dn) bn (k) = dn-l (k) -Wn ■ bnrl (k -Dn), where k = 0,l,2,...,2N-l, n=l,2,...,N; Dn=2Pn; an (k) and bn (k) are the complementary pair of sequences of length 2N ; 6( k) is the Kronecker delta function; k is an integer representing the time scale; n is the iteration number. Dn is delay, Pn,n -1,2,...,N, is any permutation of numbers {0,1,2,...,N-1}; and Wn is an arbitrary complex number of unit magnitude. 14. The method as claimed in claim 1, wherein one of the Golay complementary sequences has minimal aperiodic autocorrelation values for all non-zero delays of that one sequence and a maximal autocorrelation value for a zero delay of that one sequence. 15. The method as claimed in claim 1, wherein the base station performs a correlation process upon signals received from the mobile station using at least one of the Golay complementary sequences to detect a time in the correlation process when the a maximal autocorrelation value is obtained. 16. The method as claimed in claim 1, wherein a transmitter in the mobile station transmits at least one of the Golay complementary sequences in a preamble of random access burst over a random access channel. 17. The method as claimed in claim 1, wherein a length of each Golay complementary sequences is 4096. 18. The method as claimed in claim 1, wherein a set of 2 N orthogonal Golay complementary sequences is obtained from a common permutation vector Pin and a set of 2N-2 weighting vectors Wn, wherein the set of weighting vectors is an N-bit long binary representation of different, even positive integers or different, odd, positive integers taken from a set of integers {0,1,...,2N-1}, and wherein a zero in the binary representation is mapped to one of {- 1,+ 1} and a one in the binary representation is mapped to the other of {-1,+ 1}. 19. The method as claimed in claim 18, where ao (k) = A(k) and bo (k) = B (k) form a pair of binary sequences of length T and the delay Dn = T*2Pn. 20. The method as claimed in claim 18, further comprising: directly generating a pair of Golay complementary sequences using a sequence generator that requires only N memory elements. 21. The method as claimed in claim 18, further comprising: directly generating the pair of Golay complementary sequences using a sequence generator that includes a modulo 2N counter. 22. The method as claimed in claim 1, further comprising: correlating at the mobile station a received spread spectrum signal transmitted by the base station using one of the pair of Golay complementary sequences of length L= 2N, where N is any positive integer including: representing the one Golay sequence as a concatenation of two shorter, constituent complementary sequences A(k) and B(k) of length X, each constituent sequence having been repeated L/X times and modulated by +1 or -1 according to a signature sequence of length Y=L/X, where an order of concatenation is determined by an interleaving sequence of length Y = L/ X, whose elements are 0 or 1; using a Golay correlator of a length X receiving the spread spectrum signal to generate an intermediate pair of correlation values corresponding to the constituent Golay complementary sequences; storing the intermediate correlation values at a first correlation output of the Golay correlator in a first group of Y = (L/X) -1 or less serially concatenated delay lines, where each delay line has X memory elements; storing the intermediate correlation values at a second correlation output of the Golay correlator in a second group of Y = (L/ X) -1 or less serially concatenated delay lines, where each delay line has X memory elements; multiplying the first correlator output and the outputs of the first group of delay lines by respective elements of a reverted version of the signature sequence; summing only those multiplied intermediate correlation values from the first correlation output and attached delay lines which correspond to non- zero elements of a reverted version of the interleaving sequence, and only those multiplied intermediate correlation values from the second correlation output and attached delay lines which correspond to non-zero elements of the inverted and reverted version of the interleaving sequence, where the first element of the reverted version of the interleaving sequence corresponds to the correlator output and the last element of such sequence corresponds to the output of the last delay line in the cascade; and outputting a result of the summing step as a final correlation value. 23. The method as claimed in claim 22, wherein the received spread spectrum signal is non- coherently correlated using one of the Golay complementary sequence. 24. The method as claimed in claim 22, wherein the received spread spectrum signal is coherently correlated using one of the Golay complementary sequence, further comprising: multiplying the second correlator output and the outputs of the second group of delay lines by respective elements of a reverted version of the signature sequence. 25. Apparatus for use in synchronizing a mobile station with a base station using a primary synchronization code and plural secondary synchronization codes transmitted by the base station, comprising: receiver means for receiving the primary synchronization coded plural secondary synchronization codes transmitted by the base station; sequence generator means for providing one of a set of Golay complementary sequences as the primary synchronization code and plural other ones of the set of Golay complementary sequences with minimal cross-correlation with the primary synchronization code as the secondary synchronization codes; and synchronization means for obtaining a primary synchronization and a secondary synchronization with the base station using the provided primary and secondary synchronization codes. 26. The apparatus as claimed in claim 25, further comprising means for selecting the primary synchronization code and 17 secondary synchronization codes from a set of 256 orthogonal Golay sequences. 27. The apparatus as claimed in claim 25, wherein the minimal cross-correlation is minimal aperiodic cross-correlation. 28. The apparatus as claimed in claim 25, wherein the means for synchronizing includes a bank of Golay correlators to correlate a signal received by the mobile station with the primary and secondary synchronization codes. 29. The apparatus as claimed in claim 25, wherein the set of Golay complementary sequences has a property that the sum of their aperiodic autocorrelations equals zero for all non-zero shifts, the apparatus further comprising: a matched filter corresponding to a pair of Golay complementary sequences to correlate a received signal with at least one of the Golay complementary pair of sequences; a peak detector for detecting a peak output from the matched filter; means for generating a timing estimate using the detected peak output to synchronize transmissions between the mobile station and the base station. 30. The apparatus as claimed in claim 29, further comprising: means for using complex correlation values corresponding to detected peak outputs as initial fading channel coefficients employed to demodulate a message include in the received signal. 31. The apparatus as claimed in claim 29, wherein the matched filter is configured to correlate the received signal with both Golay complementary sequence, at the same time. 32. The apparatus as claimed in clam 29, wherein the matched filter correlates the received signal with the complementary sequences recursively. 33. The apparatus as claimed in claim 25, wherein at least one of the Golay complementary sequences is included in a preamble portion of a random access message transmitted by the mobile station to the base station over a random access channel. 34. The apparatus as claimed in claim 25, further comprising means for generating each Golay complementary sequence by concatenating a pair of shorter Golay sequences. 35. The apparatus as claimed in claim 25, wherein the sequence generator means includes means to generate a complementary pair of sequences in accordance the following: ao (k) = 5( k) bo(k)= 8 (k) an(k) = an-i(k) + Wn -bn-i (k -Dn) bn (k) = an-i (k) -Wn bn-i (K-Dn), where k=0,l,2,...,2N -1; n=l,2,...,N; Dn =2Pn; an (k) and bn (k) are the complementary pair of sequences of length 2N; S(k) is the Kronecker delta function; k is an integer representing the time scale; n is the iteration number, Dn is a delay, Pn,n=i,2,...,N, is any permutation of numbers {0,l,2,...,N-l};and Wn is an arbitrary complex number of unit magnitude. 36. The apparatus as claimed in claim 25, wherein one of the Golay complementary sequences has minimal aperiodic autocorrelation values for all non-zero delays of that one sequence and a maximal autocorrelation value for a zero delay of that one sequence. 37. The apparatus as claimed in claim 25, further comprising a transmitter in the mobile station for transmitting at least one of the Golay complementary sequences in a preamble of random access burst over a random access channel. 38. The apparatus as claimed in claim 25, wherein a length of each Golay complementary sequences is 4096. 39. The apparatus as claimed in claim 25, wherein the sequence generator means is configured to generate a set of 2N orthogonal Golay complementary sequences from a common permutation vector Pn and a set of 2N-1 weighting vectors Wn, wherein the set of weighting vectors is an N-bit long binary representation of different, even positive integers or different, odd, positive integers taken from a set of integers {0,1, ...,2N -1} , and wherein a zero in the binary representation is mapped to one of {-1, + 1} and a one in the binary representation is mapped to the other of {- 1, + 1}. 40. The apparatus as claimed in claim 39, where ao (k) = A( k) and bo (k) = B(k) form a pair of binary sequences of length T and the delay Dn = T*2pn. 41. The apparatus as claimed in claim 39, wherein the sequence generator means requires only N memory elements. 42. The apparatus as claimed in claim 39, wherein the sequence generator means includes a modulo 2N counter. 43. The apparatus as claimed in claim 26, wherein the synchronization means is configured to correlate at the mobile station a received spread spectrum signal transmitted by the base station using one of the pair of Golay complementary sequences of length L = 2N, where N is any positive integer, the synchronization means including: means for representing the one Golay sequence as a concatenation of two shorter, constituent complementary sequences A(k) and B(k) of length X, each constituent sequence having been repeated L/X times and modulated by + 1 or -1 according to a signature sequence of length Y = L/ X , where an order of concatenation is determined by an interleaving sequence of length Y = L/ X, whose elements are 0 or 1; means for using a Golay correlator of a length X receiving the spread spectrum signal to generate an intermediate pair of correlation values corresponding to the constituent Golay complementary sequences; means for storing the intermediate correlation values at a first correlation output of the Golay correlator in a first group of Y = (L/ X) - 1 or less serially concatenated delay lines, where each delay line has X memory elements; means for storing the intermediate correlation values at a second correlatibn output of the Golay correlator in a second group of Y = (L/ X) -1 or less serially concatenated delay lines, where each delay line has X memory elements; means for multiplying the first correlator output and the outputs of the first group of delay lines by respective elements of a reverted version of the signature sequence; means for summing only those multiplied intermediate correlation values from the first correlation output and attached delay lines which correspond to non-zero elements of a reverted version of the interleaving sequence, and only those multiplied intermediate correlation values from the second correlation output and attached delay lines which correspond to non-zero elements of the inverted and reverted version of the interleaving sequence, where the first element of the reverted version of the interleaving sequence corresponds to the correlator output and the last element of such sequence corresponds to the output of the last delay line in the cascade; and means for outputting a result of the summing step as a final correlation value. 44. The apparatus as claimed in claim 43, wherein the received spread spectrum signal is non- coherently correlated using one of the Golay complementary sequence. 45. The apparatus as claimed in claim 43, wherein the received spread spectrum signal is coherently correlated using one of the Golay complementary sequences the apparatus further comprising: means for multiplying the second correlator output and the outputs of the second group of delay lines by respective elements of a reverted version of the signature sequence. [JAYANTA PAL] OF REMERY & SAGAR ATTORNEY FOR THE APPLICANT[S] Dated this 3rd day of September, 2001 |
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in-pct-2001-01036-mum-assignment(12-03-1999).pdf
in-pct-2001-01036-mum-cancele pages(07-04-2005).pdf
in-pct-2001-01036-mum-claims(granted)(07-04-2005).doc
in-pct-2001-01036-mum-claims(granted)-(07-04-2005).pdf
in-pct-2001-01036-mum-correspondence(ipo)-(05-01-2007).pdf
in-pct-2001-01036-mum-correspondence1(24-03-2006).pdf
in-pct-2001-01036-mum-correspondence2(10-07-2006).pdf
in-pct-2001-01036-mum-drawing(07-04-2005).pdf
in-pct-2001-01036-mum-form 1(07-04-2005).pdf
in-pct-2001-01036-mum-form 13(26-06-2006).pdf
in-pct-2001-01036-mum-form 19(29-03-2004).pdf
in-pct-2001-01036-mum-form 2(granted)(07-04-2005).doc
in-pct-2001-01036-mum-form 2(granted)-(07-04-2005).pdf
in-pct-2001-01036-mum-form 3(31-08-2001).pdf
in-pct-2001-01036-mum-form 5(31-08-2001).pdf
in-pct-2001-01036-mum-pct-isa-210(07-04-2005).pdf
in-pct-2001-01036-mum-power of attorney(07-04-2005).pdf
in-pct-2001-01036-mum-power of attorney(09-05-2001).pdf
in-pct-2001-01036-mum-power of attorney(13-05-200).pdf
| Patent Number | 204152 | ||||||||
|---|---|---|---|---|---|---|---|---|---|
| Indian Patent Application Number | IN/PCT/2001/01036/MUM | ||||||||
| PG Journal Number | 23/2007 | ||||||||
| Publication Date | 08-Jun-2007 | ||||||||
| Grant Date | 05-Jan-2007 | ||||||||
| Date of Filing | 03-Sep-2001 | ||||||||
| Name of Patentee | TELEFONAKTIEBOLAGET LM ERICSSON [PUBL] | ||||||||
| Applicant Address | S-126 25 STOCKHOLM SWEDEN. | ||||||||
Inventors:
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| PCT International Classification Number | H04L7/00 | ||||||||
| PCT International Application Number | PCT/SEOO/00433 | ||||||||
| PCT International Filing date | 2000-03-03 | ||||||||
PCT Conventions:
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