Title of Invention

A METHOD OF COMMUNICATION USING ORTHOGONAL FREOUENCY DIVISION MULTIPLEXING (OFD M), A TRANSMITTER AND RECEIVER FOR USE THEREIN .

Abstract A method of communication using Orthogonal Frequency Division Multiplexing ("OFDM") includes a step of generating bit streams and the corresponding sets of frequency domain carrier amplitudes modulated as OFDM symbols to be transmitted. A next step includes inserting prefixes as guard intervals in said sample streams. A next step includes transmitting said OFDM symbols. A next step includes using prefix information to estimate the Channel Impulse Response of the transmission channels. A next step includes using the estimated Channel Impulse Response to demodulate said bit streams in the received signals, wherein said prefixes are deterministic and are known to both the receiver and transmitter. The prefixes can be a vector common to the symbols and multiplied by a weighting factor that is different between symbols. The weighting factor can have a complex pseudo-random value.
Full Text A METHOD OF COMMUNICATION USING ORTHOGONAL
FREQUENCY DIVISION MULTIPLEXING (OFDM),
A TRANSMITTER AND RECEIVER FOR USE THEREIN
DESCRIPTION
FIELD OF THE INVENTION
This invention relates to a method of communication using orthogonal
frequency division multiplexing (OFDM), a transmitter and receiver for use therein and,
more particularly, to channel estimation and tracking in [OFD] communication
Background of the Invention
OFDM communication has been chosen for most of the modern high-data
rate communication systems (Digital Audio Broadcast - DAB, Terrestrial Digital
Video Broadcast - DVB-T, and Broadband Radio Access Networks - BRAN such
as HIPERLAN/2, IEEE802.11a, for example). However, in most cases the receiver
needs an accurate estimate of the channel impulse response.
In many known OFDM systems, each OFDM symbol of size NN+ is
preceded by a guard interval that is longer than the channel impulse response
(CIR) and a cyclic prefix of D N+ samples is inserted as the guard interval at the
transmitter, the prefix consisting of D samples circularly replicated from the useful
OFDM symbol time domain samples. The cyclic prefix enables very simple
equalisation at the receiver, where the cyclic prefix is discarded and each
truncated block is processed, for example using Fourier Transform (usually Fast
Fourier Transform), to convert the frequency-selective channel output into N
parallel flat-faded independent sub-channel outputs, each corresponding to a
respective sub-carrier. For equalisation purposes, numerous strategies exist.
Following the zero forcing approach, for example, each sub-channel output is,
unless it is zero, divided by the estimated channel coefficient of the corresponding
sub-carrier.
Like other digital communication systems, OFDM modulation encounters
problems at high Doppler spreads, which occur notably when the user is moving

fast, for example in a car. HIPERLAN/2, for example, was designed to work only
up to speeds of 3 m/s ("pedestrian speed"). Accordingly, the channel impulse
response needs to be constantly tracked and updated, especially in the presence
of high Doppler spreads.
In a known OFDM communication system pilot tones are added which may
change their position from one OFDM symbol to another. The amplitudes and
positions of the pilot tones are known to the receiver. The receiver uses the pilot
tones to estimate the channel coefficients of the corresponding carriers. This
method is widely used, but it degrades the system performance, since a certain
number of carriers cannot be used for data, since they are reserved for the pilot
tones.
It is also known to add learning sequences (See for example EBU Review
Technical No. 224, August 1987, "Principles of modulation and channel coding for
digital broadcasting for mobile receiver", by M. Alard and R. Lassalle.). In
HIPERLAN/2, for example, there are at least 2 learning OFDM symbols per frame
(i.e. 2 OFDM symbols of 2 • 4/us duration in total per 2ms). If the channel changes
quickly, there must be many more training sequences and the consequence is an
even bigger degradation in the system performance.
Many of the known systems are unable to decode all carriers of OFDM
symbols in the presence of channel nulls. Recent innovations propose ways for
decoding OFDM symbols even in the presence of channel nulls (see for example
the publication entitled "Reduced Complexity Equalizers for Zero-Padded OFDM
transmissions" by B. Muquet, Marc de Courville, G. B. Giannakis, Z. Wang, P.
Duhamel in the proceedings of the International Conference on Acoustics Speech
and Signal Processing ('ICASSP') 2000 and the publication entitled "OFDM with
trailing zeros versus OFDM with cyclic prefix: links, comparisons and application to
the HiperLAN/2 system" by Muquet, B.; de Courville, M.; Dunamel, P.; Giannakis,
G. in the proceedings of the IEEE International Conference on Communications,
2000, Volume: 2. However, these publications do not offer responses to the
problems referred to above concerning channel estimation and channel tracking.
Ideally, the OFDM modulation system would keep all the advantages of
classic OFDM and additionally allow very simple and completely blind channel

estimation at the receiver. No additional redundancy would be added to the
system and therefore no bandwidth would be lost. Such a system would be
advantageous in low-mobility scenarios and would make OFDM systems
applicable to high-mobility scenarios as well.
Many of the examples and illustrations presented below are based on the
assumption N = 4 • D, that is to say that the size of the prefix (0 samples) is
assumed to be one quarter of the size of the useful OFDM symbol (N samples).
This corresponds to the case of HiperLAN/2 or IEEE802.11. This restriction is
introduced for sake of simplicity only. It will be appreciated that the examples and
illustrations are applicable more generally to the case of NN+, D N+, the
necessary adaptation being basically straightforward.
Summary of the invention
The present invention provides a method of, and a transmitter and a receiver
for, communication using OFDM as described in the accompanying claims.
Brief Description of the Accompanying Drawings
Figure 1 is a block schematic diagram of an OFDM communication system
comprising a transmitter and a receiver in accordance with one embodiment of the
invention, given by way of example,
Figure 2 is a schematic diagram of an OFDM frame in a signal appearing in
operation of the system of Figure 1,
Figure 3 is a matrix equation representing the channel impulse response for
inter-biock interference in operation of the system of Figure 1,
Figure 4 is a matrix equation representing the channel impulse response for
inter-symbol interference in operation of the system of Figure 1,
Figure 5 Is a matrix equation representing the combined channel impulse
response in operation of the system of Figure 1,
Figure 6 is a representation of a sub-matrix corresponding to the combined
channel impulse response in operation of the system of Figure 1 for a prefix part of
the signal of Figure 2,

Figure 7 represents the upper triangular sub-matrix of the channel matrix
presented by Figure 6,
Figure 8 represents the lower triangular sub-matrix of the channel matrix
presented by Figure 6 , and
Figure 9 is a matrix equation representing signals appearing as a result of the
combined channel impulse response in operation of one embodiment of a system
of the kind shown in Figure 1,
Figure 10 is a matrix equation representing signals appearing during channel
estimation in operation of one embodiment of a system as shown in Figure 1,
Figure 11 is a matrix equation representing signals appearing as a result of
the combined channel impulse response in operation of another embodiment of a
system of the kind shown in Figure 1
Figure 12 is a graph representing preferred values of prefixes used in the
system of Figure 1.
Detailed description of the preferred embodiments
Figure 1 shows an OFDM communication system in accordance with one
embodiment of the invention comprising a transmitter comprising an OFDM
modulator 1 and a receiver comprising an OFDM demodulator 2, the transmitter
and the receiver communicating over a communication channel 3.
An input bit-stream is modulated onto a set of N
carriers whose carrier amplitudes are given by the vector
, corresponding to OFDM symbol number k.
Afterwards, the time domain OFDM signal is generated by means 4 which
performs an Inverse Fourier Transform operation, or preferably an Inverse Fast
Fourier Transform ('IFFT) operation where (.)r
is the transposition operator and (•)* is the complex conjugate operator:



The resulting parallel signal x (k) vector is converted to a series signal by a
parallel-to-series converter 5, a prefix, represented by the Dxl vector
being inserted into the signal as guard interval between each
OFDM symbol to produce a series digital signal xn. The series digital signal xn is
then converted to an analogue signal x(t) by a digital-to-analogue converter 6 and
transmitted over the channel 3.
The channel 3 has a Channel Impulse Response H(k) = C(k) and also
introduces noise v.
At the receiver 2, an analogue signal r(t) is received and converted to a
digital signal rn by an analogue-to-digital converter 7. The digital signal rn is then
converted to a parallel signal by a series-to-parallel converter r(k) and equalised
and demodulated by equalisation and demodulation means 9 to produce
demodulated signals s"'(k). In the following analysis, consideration of noise is
omitted for the sake for simplicity. However, including the consideration of noise
does not significantly modify the results.
In some known OFDM communication systems, the guard interval is used to
add some redundancy (D samples of redundancy are added) by introducing a
cyclic prefix, for example in the following manner:

In other words, data from the end of the frame is repeated by the transmitter in the
guard interval to produce a prefix.
In accordance with this embodiment of the present invention, however, the
prefix samples inserted as guard interval of OFDM symbol number k, ak.c0 to ak.Co-
1, are deterministic and are known to said receiver as well as to said transmitter.
The prefixes comprise a vector that is common to the
symbols multiplied by at least one weighting factor ak, so that the prefixes have the
overall form The weighting factor ak may be constant from one

symbol to another. However, in a preferred embodiment of the invention, the
weighting factor ak differs from one symbol to another, the elements of a given
vector PD being multiplied by the same weighting factor. With an OFDM modulator
in the transmitter functioning in this way, blind channel estimation in the receiver
can be done simply and at low arithmetical complexity. In particular, the receiver
can constantly estimate and track the channel impulse response without any loss
of data bandwidth. Moreover, the demodulator at the receiver can have
advantageous characteristics, ranging from very low arithmetical cost (at medium
performance) to high arithmetical cost (very good system performance).
More particularly, in the preferred embodiment of the invention, the prefix of
D samples that is added in the guard interval comprises a pre-calculated suitable
vector of D samples that is independent of the data and that is
weighted by a pseudo-random factor ak that only depends on the number k of the
latest OFDM symbol:

For the purposes of the analysis below, a second prefix/OFDM symbol vector
is defined as follows:

Several choices for are possible. It is possible to choose that is to
say that can be of any complex value. However, any ■ leads to
performance degradation compared to preferred embodiments of the invention.
It is possible to limit the choice of ak, somewhat less generally to with
This choice usually leads to good system performance, but the decoding
process risks to be unnecessarily complex.
Accordingly, in the preferred embodiment of the present invention, the phase
of ak is chosen so that where m is an integer, N is the useful OFDM
symbol size and D is the size of the pseudo-random prefix. This choice is

particularly advantageous when using the specific decoding methods described
below.
For the sake of simplicity, the following analysis assumes that the weighting
factor has been chosen as m integer. However, it will be appreciated
that the mathematical adaptation to any of the cases presented above is
straightforward.
It proves to be very useful to choose ak such that its phase changes from
OFDM symbol to OFDM symbol. The constant prefix PD is preferably chosen with
respect to certain criteria, for example the following:
• In the frequency domain, PD is as flat as possible over the frequency
band used for data carriers.
• In the frequency domain, PD is as near to zero as possible for all
unused parts of the band.
• In the time domain, PD has a low peak-to-average-power-ratio
(PAPR).
• The length of PD is the size of the OFDM guard interval, that is to say
D samples. Alternatively, a shorter sequence of length D chosen where D-D zeros are appended.
With these criteria, without any complication of the transmitter, the receiver is
able to estimate the channel impulse response blindly, track the changes of the
channel impulse response blindly and perform an arithmetically simple
equalization.
An example of a frame of OFDM symbols in accordance with a preferred
embodiment of the invention is illustrated in Figure 2. The operation of the system
will first be described for the specific case where. ak is constant and equal to 1.
Now, the modulation unit of the transmitter is clearly defined. In the following,
the operations to be performed in the receiver are considered. Each received
OFDM symbol selected at the input of the demodulator 9 can then be expressed
as follows (neglecting additive noise):


where the channel impulse response of the demodulator 9 is assumed to be
is the contribution of the demodulator 9 channel matrix
corresponding to inter-block-interference and is its contribution to inter-
symbol-interference.
The components of the received signal corresponding to inter-block-
interference are illustrated in Figure 3, where blank elements
correspond to zero values, for an example where N = 4-D (for example, in the
case of HiperLAN/2 or IEEE802.11, N = 64 and D = 16). It will be seen that [HIBI]
is a matrix of size (N+D)x(N+D) with a triangular sub-matrix of size (D-1)x(D-
1) at its upper right-hand corner, illustrated by Figure 7, the other elements of the
matrix being zero.
The components of the received signal corresponding to inter-symbol-
interference are illustrated in Figure 4, for the same case and in the
same manner as Figure 3. It will be seen that is a matrix of size
with triangular sub-matrices on its major diagonal as illustrated
by Figure 7 and triangular sub-matrices of size DxD on the diagonal
immediately below the main diagonal as illustrated by Figure 8, the other elements
of the matrix being zero.
The channel impulse response seen by demodulator 9 is represented by the
sum of the inter-block-interference and the inter-symbol-interference
as shown in Figure 5. The resulting signal for this example is shown in Figure 9,
where are successive parts of the OFDM symbol #k containing as
well contributions of the preceding and following prefix convolved by the channel,
are corresponding parts of size D of the useful signal transmitted
and is a corresponding part of size D of the following prefix in this example.
Of course, the example may be generalised to any
The expectation values of the parts of the received signals are as follows:


It will be appreciated that the expectation values of the useful parts x0(k) to
x3(k) of the OFDM symbol tend to zero over a large number of symbols since they
are quasi-random with zero mean. However, the prefix PD is known to the receiver
(and in this embodiment is constant over successive symbols) and enables
to be estimated, by approximating the expectation values E0 and
E4 over a large number R of symbols:

The sum of the expectation values E0 and E4 is then given by:

A first embodiment of a method of channel impulse response estimation on D
symbols in accordance with the present invention utilises the expression of the
above equation as follows:


where the matrices are the (Fast) Fourier
Transform and Inverse (Fast) Fourier Transform matrices respectively and the
prefix PD is of size D. The matrices are illustrated by Figure 7,
Figure 8 and Figure 6 respectively.
Accordingly, in this first method, the channel impulse response is estimated
using the following steps:

• Perform a component-by-component division of the first result by the

The resulting channel estimation is hD of size Dxl. This method works well
in many circumstances and has a low arithmetic cost, since its calculations are
based on matrices of size DxD. However, an OFDM symbol which usually is of
size N>D samples will be equalized based on this estimation. Thus, this method
works very well if the prefix-spectrum is non-zero everywhere in the FFTDXD
domain (and, of course, everywhere well above channel noise). This can be a
troublesome limitation in other circumstances.
A second embodiment of a method of channel impulse response estimation
on D carriers in accordance with the present invention avoids this limitation, at the
expense of increased arithmetic cost. This second method does not estimate hD
based on a de-convolution in the FFTDXD domain as presented above, but
estimates directly based on the received vector



This equation is represented in more detail in Figure 10. In this second
method, the channel impulse response is estimated using the following steps:

The last step of the list presented above is not essential for the basic
equalization algorithm but may be useful, for example in algorithms used to reduce
noise levels.
The above methods have been described with reference to the specific case
where ak is constant and equal to 1. In preferred embodiments of the invention,
however, the weight ak of the prefix to each symbol k is a preferably complex
pseudo-random factor that only depends on the number k of the latest OFDM
symbol. The adaptations to this method of the basic equations (shown in Figure 9)
are shown in Figure 11.
It is found that equations 4 and 8 are to be adapted as follows:

The procedures for blind channel estimation described above remain
applicable by setting This amounts to weighting the
preceding and following D prefix-samples of each received symbol by the
corresponding respectively.
The values of the prefixes are chosen as a function of selected
criteria, as mentioned above. Values that have been found to give good results
with the criteria:

• Low Peak-to-Average-Power-Ratio of the time domain signal
• Low Out-of-Band Radiation, that is to say maximise the energy of the prefix
over the useful band and not waste prefix energy over null carriers

• Spectral Flatness, e.g. SNR of each channel estimates shall be approx.
constant
• Low-Complexity Channel Estimation, i.e. by prefix spectrum whose spectral
contributions are mainly just phases (i.e. of constant modulus),
are shown in Figure 12 by way of example, for the following OFDM parameters:
• Size of the Prefix in Time Domain: 0=16 Samples
• Size of the OFDM symbols in the frame: W=64 Samples
• Carriers where channel coefficients are to be estimated (over N+D=80
carriers): Carriers 1 to 52
• Out-of-Band region: Carriers 76 to 80
• Maximum PAPR has not been limited
• Out-of-Band Radiation as low as possible
• Spectral Flatness as good as possible.
The channel estimation is done by calculating the expectation value over a
number of samples of the received vector as explained above. If the tracking of the
channel is done based on a first estimation of the channel
impulse response and a number R of OFDM symbols, the first estimate is then
updated as follows:

based on the ideas of the first method for channel estimation that has been
presented above. Alternatively, the second method can be applied by


where the factors are positive real numbers that are used for
normalization and weighting of the different contributions. Thus, for example it is
possible to take older OFDM symbols less into account for the channel estimation
than later ones. The Fourier matrix [F] can be chosen in the N+D carriers or D
carriers domain
Several equalization methods are advantageous using the pseudo-random
prefix OFDM. In general, the different methods offer different performance-
complexity trade-offs.
A first embodiment of a method of equalization uses zero forcing in the N+D
Domain and offers low complexity equalization.
With the Channel Impulse Response matrix can be represented
as follows:

Still assuming that the length of the channel impulse response is D, the
coefficients are set to zero. This is a so-called pseudo-circulant
matrix, corresponding to the case where is not equal to 1, and can be
diagonalized as follows:


With the assumption that and that the weighting factor
has been chosen as , where m is an integer, and if the received
vector R(k) is:

the procedure of this method of zero forcing equalisation is:
• Perform Multiplication

• Calculate the frequency shifted, estimated CIR coefficients

• Extract the N equalized samples of the kih OFDM-data symbol to SEQ(k).
• Transform the kth OFDM data symbol. into the frequency domain by
a Fourier transform
• Proceed with metric calculation, etc. on the received equalised carriers.
Another embodiment of a method of equalization uses a method known from
studies on zero padding. The received vector R(k) in the OFDM Pseudo Random
Prefix Scheme can be expressed as follows, where [P] contains a (N+D)xN pre-
coding matrix and IN is the NxN identity matrix:

The Channel Impulse Response estimation obtained as described above is
then used together with the known values of PD to perform the following operation
SUBSTITUTE SHEET (RULE 26)


in which the known prefix values are multiplied by the Channel Impulse Response
estimation and the result subtracted from the received signal. In the general case,
[H\ is a pseudo circulant channel matrix. So, the diagonalisation of such matrices
can then be performed in order to calculate [H]PD efficiently. Then, several
equalization approaches are possible, for example the Zero Forcing (ZF) approach
or the Minimum Mean Square Error (MMSE) equalization approach. Examples of
MMSE equalization methods are described in the articles "OFDM with trailing
zeros versus OFDM with cyclic prefix: links, comparisons and application to the
HiperLAN/2 system" by Muquet, B.; de Courville, M.; Dunamel, P.; Giannakis, G.
ICC 2000 - IEEE International Conference on Communications, Volume 2, 2000
and "Reduced Complexity Equalizers for Zero-Padded OFDM transmissions" by
Muquet, B.; de Courville, M.; Giannakis, G. B.; Wang, Z.; Duhamel, P.
International Conference on Acoustics Speech and Signal Processing (ICASSP)
2000.
In one example, the equalisation is performed based on a zero-forcing
approach by multiplying ym by the Moore-Penrose pseudo-inverse [G] of the


The definition of the Moore-Penrose pseudo-inverse is, among others,
discussed by Haykin in the book: "Adaptive Filter Theory" by Simon Haykin, 3rd
edition, Prentice Hall Information and System Science Series, 1996. Haykin uses
the common definition


where [A] is a rectangular matrix.

WE CLAIM :
1. A method of communication using Orthogonal Frequency Division Multiplexing
("OFDM"), the method comprising the steps of:
generating bit streams and the corresponding sets of
frequency domain carrier amplitudes where k is the OFDM symbol
number, modulated as OFDM symbols to be transmitted from a transmitter,
inserting prefixes as guard intervals in said sample streams,
transmitting said OFDM symbols from said transmitter to a receiver,
using information from said prefixes to estimate the Channel Impulse Response of
the transmission channels at the receiver, and
using the estimated Channel Impulse Response to demodulate said bit streams in the
signals received at said receiver, wherein said prefixes are
deterministic and are known to said receiver as well as to said transmitter.
2. A method of communication as claimed in claim 1, wherein said prefixes
comprise a vector that is common to said symbols multiplied by at least one
weighting factor
3. A method of communication as claimed in claim 2, wherein said weighting factor
differs from one symbol to another but the elements of a given vector are multiplied by the
same weighting factor.
4. A method of communication as claimed in claim 3, wherein said weighting factor
has a pseudo-random value.

5. A method of communication as claimed in claim 1, wherein said weighting factor is
a complex value.
6. A method of communication as claimed in claim 5, wherein the modulus of said weighting
factor is constant from one symbol to another.
7. A method of communication as claimed in claim 6, wherein said weighting factor is
proportional to where N is the useful OFDM symbol size, D is the size of the prefix
vector and m is an integer.
8. A method of communication as claimed in claim 1, wherein estimating said Channel
Impulse Response comprises performing a Fourier Transform on a first vector
that comprises the received signal components corresponding to one of said prefixes
and also the received signal components corresponding to the
following one of said prefixes to produce a
received prefix signal transform performing a similar Fourier Transform on a second
vector (Vp) that comprises the known values of corresponding components of said prefixes
to produce a known prefix transform , and
performing a component-by-component division of the receiving prefix, signal transform
by known prefix transform

9. A method of communication as claimed in claim 8, wherein said prefixes comprise a
vector that is common to said symbols multiplied by weighting factors , said
weighting factors differing from one symbol to another but the elements of a given vector being
multiplied by the same weighting factor, and wherein the received signal components
corresponding to said one of said prefixes and said following one of said
prefixes are weighted by the respective value of said weighting factor
before summing and performing said Fourier Transform to produce said received
prefix signal transform
10. A method of communication as claimed in claim 8, wherein said Fourier Transforms are
of dimension DxD, where D is the size of said prefixes
11. A method of communication as claimed in claim 8, wherein said Fourier Transforms are
of dimension where D is the size of said prefixes and N is
thesize of the OFDM signals between said prefixes, said first vector comprises said sum of
said received signal components corresponding to one of said prefixes and of
the following one of said prefixes augmented by a zero value vector
of size (N) to produce said received prefix signal transform of size (N+D), and
said second vector comprises said known components of said prefixes I
augmented by said zero value vector of size i to

produce said known prefix transform of size (N+D).

12. A method of communication as claimed in claim 1, wherein estimating said Channel
Impulse Response comprises combining information from said prefixes
for more than one symbol to obtain said estimated Channel
Impulse Response
13. A method of communication as claimed in claim 1, wherein demodulating said bit
streams comprises:
performing the multiplication by a matrix proportional to

calculating the frequency shifted CIR coefficients

extracting the N equalized samples corresponding to the data symbol to the vector
and
transforming the symbol into frequency domain by performing a Fourier Transform:


14. A method of communication s claimed in claim 1, wherein demodulating said bit streams
includes padding the received signal matrix and the operator matrices with zeros to obtain
compatible dimensions for subsequent operations, multiplying the known prefix value matrix by
the Channel Impulse Response estimation matrix and subtracting the result from the received
signal matrix.
15. A transmitter for use in a method of communication as claimed in claim 1 and comprising
generating means for generating bit streams modulated as OFDM
symbols to be transmitted and inserting prefixes as guard intervals between said OFDM symbols,
said prefixes being deterministic and suitable to be known to said receiver as
well as to said transmitter.
16. A receiver for use in a method of communication as claimed in claim 1 and comprising
demodulating means for receiving signals that comprise bit streams
modulated as OFDM symbols to be transmitted from a transmitter,
with prefixes inserted in guard intervals between said OFDM symbols, said OFDM symbols
having been transmitted from said transmitter to said receiver, said demodulating means being
arranged to use information from said prefixes to estimate the Channel Impulse Response
of the transmission channels and to use the estimated Channel Impulse Response
to demodulate said bit streams in the signals received at said receiver, said prefixes
being deterministic and being known to said receiver as well as to said
transmitter.

A method of communication using Orthogonal Frequency Division Multiplexing
("OFDM") includes a step of generating bit streams and the corresponding sets of
frequency domain carrier amplitudes modulated as OFDM symbols to be transmitted. A
next step includes inserting prefixes as guard intervals in said sample streams. A next
step includes transmitting said OFDM symbols. A next step includes using prefix
information to estimate the Channel Impulse Response of the transmission channels. A
next step includes using the estimated Channel Impulse Response to demodulate said
bit streams in the received signals, wherein said prefixes are deterministic and are
known to both the receiver and transmitter. The prefixes can be a vector common to the
symbols and multiplied by a weighting factor that is different between symbols. The
weighting factor can have a complex pseudo-random value.

Documents:

227330-CORRESPONDENCE 1.1.pdf

227330-PA.pdf

563-KOLNP-2005-(29-03-2012)-ASSIGNMENT.pdf

563-KOLNP-2005-(29-03-2012)-CERTIFIED COPIES(OTHER COUNTRIES).pdf

563-KOLNP-2005-(29-03-2012)-CORRESPONDENCE.pdf

563-KOLNP-2005-(29-03-2012)-FORM-16.pdf

563-KOLNP-2005-(29-03-2012)-PA-CERTIFIED COPIES.pdf

563-KOLNP-2005-FORM-27.pdf

563-kolnp-2005-granted-abstract.pdf

563-kolnp-2005-granted-claims.pdf

563-kolnp-2005-granted-correspondence.pdf

563-kolnp-2005-granted-description (complete).pdf

563-kolnp-2005-granted-drawings.pdf

563-kolnp-2005-granted-examination report.pdf

563-kolnp-2005-granted-form 1.pdf

563-kolnp-2005-granted-form 18.pdf

563-kolnp-2005-granted-form 3.pdf

563-kolnp-2005-granted-form 5.pdf

563-kolnp-2005-granted-pa.pdf

563-kolnp-2005-granted-reply to examination report.pdf

563-kolnp-2005-granted-specification.pdf

563-KOLNP-2005-OTHER PATENT DOCUMENT.pdf


Patent Number 227330
Indian Patent Application Number 563/KOLNP/2005
PG Journal Number 02/2009
Publication Date 09-Jan-2009
Grant Date 06-Jan-2009
Date of Filing 04-Apr-2005
Name of Patentee MOTOROLA, INC.
Applicant Address 1303 EAST ALGONQUIN ROAD SCHAUMBURG, IL
Inventors:
# Inventor's Name Inventor's Address
1 MUCK, MARKUS 67-69, RUDE DE LA COLONIE, F-75013, PARIS
2 DE COURVILLE, MARC 4, 43 RUE LA COLONIE, F-75013, PARIS
3 DEBBAH, MEROUANE 72, RUE CAMILLE DESMOULINS, F-94230 CACHAN
PCT International Classification Number H04L 52/02
PCT International Application Number PCT/EP2003/050767
PCT International Filing date 2003-10-30
PCT Conventions:
# PCT Application Number Date of Convention Priority Country
1 02292730.5 2002-10-31 EUROPEAN UNION